In a Doppler radar system, pulses are transmitted and echoes are received from radar-reflecting objects within range of the radar system. The echo pulses received are demodulated and in-phase and quadrature (I & Q) signals are derived. The I & Q signals are duplicate copies of the demodulated signal with a phase difference of exactly 90° between them. The I & Q signals form a complex-number phasor that is processed using a Fourier transform to obtain a Doppler spectrum of the echo signal. In the Doppler spectrum, closing and receding targets correspond to positive and negative Doppler frequencies, respectively. (A receiver system that does not produce I & Q components is typically unable to distinguish between positive and negative frequencies, and is therefore unable to determine if the target is approaching or receding.)
The I & Q demodulation is conventionally performed using an arrangement such as that shown in FIG. 1. In this arrangement 10, the received signal 20, modulated onto a carrier wave, is split between a first mixer 30 and a second mixer 40. At the first mixer 30, the signal 20 is mixed with a local oscillator signal 50 from a local oscillator at the frequency of the carrier wave. At the second mixer 40, the signal 20 is mixed with the local oscillator signal from the local oscillator 50, but before the mixing a 90 degree phase shift is introduced into the local oscillator signal. The outputs from the first and second mixers 30, 40 are filtered by first and second low-pass filters 70, 80, respectively, to produce a baseband in-phase signal 90 from the first mixer 30 and first low-pass filter 70 and a baseband quadrature signal 100 from the second mixer 40 and the second low-pass filter 80. Thus, the signal 20 is split into two pathways, one of which is multiplied by a sine wave in a first superheterodyne mixing stage at the first mixer 30, and the other of which is multiplied by a cosine wave in a second superheterodyne mixing stage at the second mixer 40. This arrangement thus translates the carrier frequency to base-band (i.e. 0 Hz carrier signal), and achieves the required 90° phase shift between the two pathways.
However, the arrangement of FIG. 1 can suffer from problems in practice, because, for proper operation, the two receiver pathways need to be closely matched in gain and phase. Any mismatch in gain or phase between the channels will result in a spurious image of the target at a frequency which is the negative of the target frequency. The need for close match in gain and phase usually results in a need for the receiver to undergo a calibration process before it can be used. Often that calibration is not completely successful, for example because the gain and phase imbalance changes with temperature and so the calibration becomes invalid as the system heats up.
A further disadvantage of the arrangement of FIG. 1 is that it requires a significant number of components.
In a traditional monopulse angle-of-arrival measurement system, the radar receiver produces a sum channel and two difference channels (azimuth and elevation in an airborne radar, or elevation and traverse in a ground-based radar). In conventional monopulse radar systems, the sum and difference of four antenna feed apertures at the output port of the radar antenna are provided by a large, heavy and relatively expensive microwave comparator unit (a summing and differencing junction implemented in a waveguide).
In phase-coded pulse compression, transmit pulses are phase-coded to allow pulse compression in the radar receiver. Phase coding of radar pulses is a well-known technique for achieving high resolution while at the same time retaining adequate signal-to-noise ratio. The operation of the technique is illustrated in FIG. 2. The transmitted RF pulse 110 has its phase switched between 0° and 180° in a randomised pattern 120 (FIG. 2(a)), resulting in an RF pulse with an applied phase code. In the receiver the carrier signal is stripped off leaving the originally applied code sequence (FIG. 2(b)), i.e. a demodulated baseband pulse. This code sequence is then compared with the phase code applied to the transmit pulse 110, in a correlator device which calculates the cross-correlation function of the transmitted and received pulses, producing a compressed pulse 130 (FIG. 2(c)) with a resolution equal to the duration of one of the phase code digits.
Lee K. Patton, in “A GNU Radio Based Software-Defined Radar”, 9 Apr. 2007 (see http://rave.ohiolink.edu/etdc/view?acc_num=wright1176142845) describes a software defined radio that can be used to create a plurality of different radar systems. At page 6 it describes a software-defined radio including (i) a receiver in which a received signal is converted from its carrier frequency to an intermediate frequency or to baseband and (ii) a transmitter in which a transmit signal is converted from an intermediate frequency or baseband to the desired carrier frequency. In both the receiver and the transmitter, an analogue-to-digital converter or digital-to-analogue converter is said to need only to convert the signal over its modulation bandwidth, and not the entire bandwidth from DC to carrier, if the signal is at baseband. At the intermediate frequency, the system must convert the signal from DC to the intermediate frequency plus the upper half of the modulation bandwidth, although Patton says that even in this case a lower rate ADC/DAC can be used.
US 2003/020653A1 (Baugh et al.) describes a system and method for narrowband pre-detection signal processing for passive coherent location applications.
EP2131209A1 (Saab AB) describes a radar receiver for processing arbitrary waveforms, in particular from a noise radar. The document is concerned with how to apply the double spectral processing used in noise radar systems to a wideband digital radar system. A waveform generator generates an arbitrary noise waveform having a flattened frequency spectrum. An undersampling analogue-to-digital converter is used to fold back the wide frequency band of the analogue wide-band radar return waveform into the baseband of said converter. Spectral processing is performed on the power spectrum of the undersampled digital wide-band waveform in order to obtain a discrete ripple frequency power spectrum. Ripple frequencies indicating radar targets are located in the discrete ripple frequency power spectrum. The ripple frequencies are said to remain basically unaffected by the aliasing caused by the undersampling, and therefore to be identifiable in the discrete ripple frequency power spectrum of the undersampled digital radar waveform.
US 2002/012200A1 (Bradley et al.) describes a ground penetrating radar system, including an RF module and a digital module.
It would be advantageous to provide a Doppler radar receiver in which one or more of the aforementioned disadvantages is eliminated or at least reduced.